The reduced cost and continued improvement in the performance of LEDs has led to their increased application in recent years. They are widely employed, for example, as illumination elements in backlighting applications, such as within the backlight of liquid crystal displays (LCDs). Backlights of this type are used to provide uniform and constant illumination of an array of LCD elements which make up the display. LEDs are also commonly employed in other applications such as within lighting assemblies, status indicators and displays on a variety of equipment and installations. Within all of these applications, LEDs are typically arranged in series connected strings and are provided with a substantially constant current, via a constant current driver circuit. Such driver circuits therefore include a current regulation means.
It is well known that variations in the drive current supplied to an LED, or a chain of LEDs, forming part of a lighting system can adversely affect the performance of the system. For example, in large lighting or signage applications, uncertainty in the drive current can lead to corresponding uncertainty in power consumption. Such uncertainties are generally unwelcome in the context of a lighting technology marketed on the basis of energy conservation. As well as this, variations in current can, in certain applications requiring, for instance, Red-Green-Blue (RGB) colour mixing, result in variations in the chromatic properties of an illuminated platform, such as a sign. Furthermore, the useful lifetime of an LED, or series-connected chain of LEDs is related to the junction temperature of the/each LED, which is in turn partly related to the current flowing through the/each LED. Therefore, precise control of LED current can result in improvements in the predictability of LED lifetime. It is further known that variations in the current supplied by an LED driver can occur as a result of variations in component properties due to either manufacturing variations, or as a result of variations in temperature. Other performance requirements for LED drivers for lighting systems, relate to the reliability of a driver. Typically, this is expressed through the use of a metric referred to as Mean Time Between Failures (MTBF). For a given electronic assembly, using well-established components, this metric can readily be calculated, provided that the electrical and thermal stresses placed on each component during operation are known. Due to the mix of components typically used in conventional so-called switch-mode LED drivers, which includes switching Metal Oxide Semiconductor Field Effect Transistors (MOSFETs) and Electrolytic Capacitors, both of which are known to have limitations in terms of long-term reliability, corresponding limitations are placed on the MTBF of such drivers. Conversely, drivers using linear means of current regulation, in place of switch-mode means, typically suffer from variations in current, referred to previously.
It is therefore highly desirable that an LED or a chain of LEDs is supplied with a substantially constant drive current. It is particularly desirable that a substantially constant drive current is produced through the use of high MTBF electronic assemblies, which use high-reliability components such as bipolar transistors and which avoid or at least limit the need for Electrolytic Capacitors. In the case of switch-mode LED drivers, wherein the current regulation function is provided by a switching voltage waveform that successively charges and discharges a circuit element such as an inductor, with such discharge taking place through an LED chain, a substantially constant current can be produced within the LED chain. The current delivered to the LED chain by such a switch-mode driver is dependent on a number of factors, including the proportion of time that the switching voltage is in the ‘ON’ state, during which it is delivering charge to the LED chain (this proportion being referred to as the Duty Cycle of the switching waveform). This switching process, however, leads to the generation of Electro-Magnetic Interference (EMI) waveforms which necessitate the use of EMI filtering structures, which in turn use Electrolytic Capacitors. From the perspective of seeking to maximize the MTBF of a driver, therefore, it can be advantageous to construct a constant current LED driver, based upon a current regulation circuit that does not use any switch-mode elements, so long as current accuracy can be maintained, including the constancy of current over temperature. The present invention is concerned with the general aim of providing a regulated current from an input voltage in order to provide a stable or substantially constant drive current for supply to illumination devices such as LEDs, or other devices which are adversely affected by, or sensitive to, current fluctuations. Preferred embodiments of the present invention seek to achieve this aim preferably without the use of switch-mode circuitry within the current regulator, thereby tending to increase the long-term reliability of the regulator, as well as reducing or eliminating the need for Electrolytic Capacitors in an LED driver based upon the regulator, thereby increasing further, the long-term reliability of the LED driver.
Current regulator devices or circuits which seek to provide a current to an LED or LED chain that is regulated, or substantially constant, with respect to supply voltage are known. So-called “constant current regulators” can be realised in either two-terminal or three-terminal topologies. FIG. 1a illustrates the case of a two-terminal regulator, whilst FIG. 1b shows a three-terminal current regulator.
However, even with the use of a current regulator device, variations in the drive current supplied to an LED chain can still arise for a number of reasons. Manufacturing spreads—i.e. variations in the manufacturing tolerance of current determining circuit elements—is one of the main causes of variations arising in the LED drive/supply current. Variations also arise due to the “temperature coefficient” of the current regulator circuit—in other words the dependence of the regulator performance with respect to ambient or junction temperature.
As will become apparent from the following discussion relating to previously considered constant current regulators, there are a number of drawbacks associated with the prior art.
FIG. 2 shows a schematic for a typical three-terminal current regulator used for the purpose of driving a chain of LEDs (also cited in US2010/0277091—Brieda et al). The minimum ‘drop voltage’ across a current regulator according to the design shown in FIG. 2 is around 1.3V—this being equal to two Base-Emitter voltage (vbe) drops (across transistors Q1 and Q2). One of these ‘vbe drops’—namely the one across the base-emitter junction of Q1—occurs across R1, resulting in a current through R1 of vbe1/R1. Assuming that Q2 is drawing negligible base current, the current through the LEDs is also equal to vbe1/R1, where vbe1 is the base-emitter voltage of transistor Q1. Consequently, due to the inherent temperature dependence of vbe, the temperature-related variation of the LED current, expressed as a fraction of nominal LED current, is given by:TC=(δILED/δT)/ILED=(δvbe1/δT)/vbe1nom  equation 1
Wherein, vbe1nom is the nominal value of vbe1 at a standard temperature (300K). In the design of FIG. 2, vbe1nom is around 0.6V and δvbe1/δT is, to a very good engineering approximation, −2 mV/K. Consequently, the lowest achievable value of the temperature coefficient, TC, for this design is −0.0033 K−1 (−0.33% per Kelvin, or −3,300 ppm per Kelvin). The currents shown for this ‘standard solution’ in Table 1 of Brieda et al indicate a variation of −0.35% per Kelvin. This value of TC would result in the current provided to the LED string varying by −/+9.25% over a temperature range of +/−55 Kelvin.
The solution proposed by Brieda et al suffers from a temperature coefficient TC of −0.0650% per Kelvin (−650 ppm/K). This results in a variation in LED current of −/+3.6% over +/−55 Kelvin. This variation renders the Brieda solution unsuitable for many applications where fluctuations in ambient temperature are expected and where the optical output, in terms of Luminous Flux and/or chromatic indices, of an assembly of LEDs is/are required to remain substantially constant.
In summary, therefore, although the Brieda design offers some advantages in terms of cost-efficiency, this design is capable of delivering minimum values of temperature coefficient, TC, of around 650 ppm/K in magnitude. This magnitude of TC is still significant and leads to variations of around −/+4% in LED current over the specified temperature range of −30 C to +80 C.
Also known in the art is a generalised two-terminal circuit topology capable of providing a substantially constant current, limited by the current and voltage handling capabilities of a Silicon bipolar transistor. This generalised topology is shown in FIG. 3.
Within this topology, a Voltage Regulating Device (VRD) is used to regulate the voltage across a series combination of a base-emitter voltage, vbe, and a current programming resistor, R. If the regulated voltage across the VRD is Vreg, then the current through the resistor R is given by:IR=(Vreg−vbe)/R  equation 2
By allowing two such currents to mutually bias the base-emitter junctions of the two bipolar transistors shown in FIG. 3, the total regulated current through the regulator is given by:IT=2·IR=2·(Vreg−vbe)/R  equation 3
The temperature coefficient of this current, defined (as before) as the fractional change in IT with temperature, is given by:TC=(δIT/δT)/IT=(δVreg/δT−δvbe/δT)/(Vreg−vbe)  equation 4
It is known in the art that for a Silicon bipolar transistor, the value of δvbe/δT is around −2 mV/K and that vbe, being the voltage across a forward-biased Silicon pn junction is around 0.7V.
The thermal behaviour of the regulated current therefore depends upon the nature and thermal behaviour of the VRD. In light of this, a particular design, based on this generalised topology has been disclosed in which the VRD comprises a series combination of a forward biased PN junction diode and a ‘bandgap reference’ diode. This design is shown in FIG. 4. For this design, the regulation voltage, Vreg is given by:Vreg=Vdiode+Vbg  equation 5
It is a property of a bandgap reference diode, that the voltage across it, Vbg (typically 1.23V) is substantially invariant with temperature, whereas, the voltage across a forward-biased PN junction diode, Vdiode, will vary with temperature in the same way as a base-emitter junction (it also being a forward-biased PN junction, carrying substantially the same current as the diode). Therefore, the thermal behavior of Vreg will be identical to that of vbe, thereby producing a zero temperature coefficient, TC, for the regulator current.
There are, however, limitations placed on the performance and cost of regulators of this design. In particular, a Silicon bandgap reference diode, maintaining a temperature stabilised voltage across it of 1.23V, operates up to a typical maximum current of 20 mA. This places an upper limit on the total regulator current, IT, of 40 mA.
Furthermore, the very low differential impedance of the bandgap diode (typically less than 1Ω) makes it difficult to ensure that devices of this type can be connected in parallel, whilst sharing current between them. FIG. 5 illustrates the problem. It depicts the I/V characteristics of two bandgap diodes, lying (for worst-case illustration) at each end of the manufacturing spread in Vbg—for a typical Silicon bandgap diode, this spread (Vbg2−Vbg1) is around 8 mV. It can readily be seen, that if two such diodes are placed in parallel, the diode with the lowest value of Vbg (Vbg1) will take a certain amount of current (shown as Ibg1) before the other diode begins to take any current. Consequently, there will be a range of VRD current, over which no current-sharing takes place and over which therefore, the current handling capabilities of the VRD and therefore of the current regulator as a whole, remains limited by the current-handling capabilities of a single bandgap reference diode.
By inspecting the I/V characteristic of a bandgap diode with a maximum current handling capability of 20 mA (such as the LT1004-1.2) it can be seen, that the voltage across Bandgap Diode 1 in FIG. 5, has a value which is substantially 8 mV higher than its nominal (low current) value, thereby ensuring that Bandgap Diode 2 is turned-on, when the current through Bandgap Diode 1 has reached a value of around 14 mA. This means that Bandgap Diode 1 and Bandgap Diode 2 do not share current, until the current through Bandgap Diode 1 has reached a value that is only a few milliamps short of its maximum rated value. Furthermore, due to the nonlinear nature of the I/V characteristic of a bandgap diode, where the differential impedance (rate of change of voltage with current) is significantly higher at low current than at high current, as the current through Bandgap Diode 1 increases by 6 mA, up to its rated maximum of 20 mA, the current through Bandgap Diode 2 will increase by significantly less than this (around 3 mA).
Consequently, replacing the bandgap diode in each VRD of a circuit according to FIG. 4, with a parallel combination of two such bandgap diodes, allowing for manufacturing variations in Vbg, can be reliably expected to increase the current handling capability of each VRD by only 9 mA, compared with the desired 20 mA. Therefore, the reliably expected increase in the current handling capability of the current regulator as a whole would be only around 18 mA, as opposed to the desired 40 mA. This is effectively a process of diminishing returns in terms of current handling per unit cost. The importance of this is significant, in view of the fact that bandgap reference diodes are not simple diode structures, but fairly complex integrated circuits, containing several circuit elements. A typical 1.23 Volt bandgap reference diode contains around 13 bipolar transistors and 8 resistors, making it a significant contributor to the overall cost of the current regulator.
An alternative approach, in the case of a circuit according to the design of FIG. 4, would be to form parallel combinations of the entire low current VRD (where each such low current VRD is, as shown, a series combination of forward-biased PN junction diode and bandgap reference diode) to form a high current VRD. This, however, would mean replicating both the bandgap diode and the PN junction diode, thereby again, increasing significantly, the cost of the regulator.
As such, the realisation of the general topology shown in FIG. 4 does not offer a cost-effective solution to the challenge of providing a low temperature coefficient current regulator which is programmable over a wide range of constant currents.